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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 1, JANUARY 2008 251 Torque Ripple and EMI Noise Minimization in PMSM Using Active Filter Topology and Field-Oriented Control Kayhan Gulez, Member, IEEE, Ali Ahmed Adam, and Halit Pastaci Abstract—This paper proposes an active filter (AF) topology to reduce the torque ripple and harmonic noises in a permanent-magnet synchronous motor. The topology consists of an insulated-gate bipolar transistor AF and two resistance– inductance–capacitance high-pass electromagnetic interference (EMI) noise filters, i.e., one in the primary and the other in the secondary circuit of the coupling 1 : 1 transformer. The AF is characterized by detecting the harmonics in the motor phase voltages by comparing the measured phase values with the ref- erence voltages generated as a function of the motor parameters and control setting values under field-oriented control. The AF uses the hysteresis voltage control method, while the motor main circuit uses the hysteresis current control method; thus, the two control methods independently work together to provide an almost sinusoidal voltage to the motor windings. The simulation results show total harmonic distortion drops of greater than 13% with EMI noise damping down to ∼−10 dB as well as considerable reduction in torque ripple. Index Terms—Active power filter, electromagnetic interference (EMI), permanent-magnet synchronous motor (PMSM), rotor reference frame, torque ripple, voltage harmonic. I. INTRODUCTION V ECTOR control with hysteresis controllers provides avery simple method to control the speed and torque of a permanent-magnet synchronous motor (PMSM). It is desirable that the voltages and currents provided to the motor terminals do not include harmonic components; in other words, they are perfect sine waves. However, various harmonic components are contained in the motor windings due to many causes, such as structural imperfectness of the motor, harmonics in the control system associated with measurement noises, switching harmon- ics, and harmonic voltages supplied by the power inverter, which constitute the major source of unavoidable harmonics in PMSM especially when the sampling period is greater than 100 µs. These harmonics cause many unwanted phenomena, such as electromagnetic interference (EMI) noise, which affects the motor control system and the torque ripple, which then provides mechanical vibrations and acoustic noise. Manuscript received March 3, 2005; revised September 21, 2007. K. Gulez and H. Pastaci are with the Department of Electrical Engineering, Electrical–Electronic Faculty, Yildiz Technical University, Istanbul 34349, Turkey (e-mail: gulez@yildiz.edu.tr; hpastaci@yildiz.edu.tr). A. A. Adam was with the Department of Electrical Engineering, Electrical–Electronic Faculty, Yildiz Technical University, Istanbul 34349, Turkey. He is now with the Faculty of Engineering Science, Omdurman Islamic University, Omdurman, Sudan (e-mail: aliadam999@yahoo.com). Digital Object Identifier 10.1109/TIE.2007.896295 Recently, many researchers have tried to reduce the torque ripple and harmonics in PMSM. Yilmaz et al. [1] presented an inverter output filter topology for pulsewidth modulation (PWM) motor drives to reduce the harmonics of PMSM. The proposed filter by Yilmaz et al. is composed of a conventional RLC filter cascaded with an LC trap filter tuned to the in- verter line frequency. The scheme shows some effectiveness in reducing the switching harmonics; however, it requires tuning to adjust the trap filter for the switching frequency, and that the voltage harmonics are still high. Harrori et al. [2] and Kim et al. [3] have proposed a suppression control method of the motor frame vibration and the rotational speed vibration of PMSM by utilizing feedforward compensation control with a generation of compensation signals to suppress the harmonic contents in the d−q control signals by repetitive control and Fourier transform. However, their work has nothing to do with the switching harmonics and voltage harmonics provided by the PWM inverter that supplies the motor. Many researchers [4]–[6] have addressed the active filter (AF) design to re- duce or compensate harmonics in the supply side by injecting harmonics into the line current, which has no effect on the current supplying the load. Degober et al. [7] have proposed an approach to minimize the torque ripple of the surface- mounted PMSM caused by back electromotive force (EMF) harmonics. The approach used self-tuning multiple-frequency resonant controllers in the Concordia reference frame with good results. However, the coefficients of the resonant controller should be reevaluated according to the rotor speed while the motor operates, and the excitation current waveforms should be predetermined according to the commanded torque and rotor position. Stamenkovic et al. [8] have provided a model that can identify the torque ripple experienced with PMSM based on measurement performed on a typical PMSM. They provided results suitable for designing active and passive torque ripple compensation. Gasc et al. [9] have proposed an approach to reduce the ripple torque without position sensor. The scheme utilizes a reduced-order torque observer and a Kalman filter. The scheme provided good results. However, accurate speed, currents, and line voltages are necessary to define the position and load torque for the observer operations. Yun et al. [10] have proposed a variable step-size normalized iterative learn- ing control (VSS-NILC) scheme to reduce the periodic torque ripple. The VSS-NILC is combined with the existing proportional-integral current controller to minimize the mean square torque error. The provided simulation results show some improvements in minimizing the torque ripple. In [11]–[19], a 0278-0046/$25.00 © 2008 IEEE 252 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 1, JANUARY 2008 modification in the control algorithm and/or voltage modulation has been carried to overcome the torque ripple problem. In [11] and [12], complicated and expensive multilevel inverters have been used to reduce the torque ripple and the fixing switching frequency. In [13], a smooth torque has been obtained by repetitive control techniques to the current control in a field-oriented PMSM drive. In [14], a method to reduce the commutation torque ripple in a sensorless motor drive has been developed. The developed method measures the commutation interval from the terminal voltage of the motor and calculates the required PWM duty ratio to suppress the commutation torque ripple. In [15], switching techniques, which insert zero- voltage vectors and/or more nonzero-voltage vectors to the conventional switching table, for ac drives with direct torque control, have been achieved. In [16], an approach called con- duction angle control has been proposed to minimize the torque ripple in a doubly salient PMSM. In [17] and [18], space vector modulation has been used to reduce the torque ripple with good results. In [19], a new direct torque control algorithm for the in- terior PMSM has been proposed to improve the performance of hysteresis direct torque control (HDTC). The algorithm uses the output of the two hysteresis controllers used in the traditional HDTC to determine two adjacent active vectors. It also uses the magnitude of the torque error and stator flux linkage position to select the switching time from the suggested table structure to reduce the complexity of calculation. The simulation results show adequate dynamic torque performance and considerable torque ripple reduction. However, the hardware requirement is relatively expensive. In this paper, we propose a filter topology to reduce the torque ripple and harmonic noises in a PMSM controlled by field-oriented control (FOC) with current hysteresis controllers. The filter topology consists of an insulated-gate bipolar tran- sistor AF and two RLC filters, i.e., one in the primary circuit and the other in the secondary circuit of a coupling transformer. The AF is characterized by detecting the harmonics in the motor phase voltages and uses the hysteresis voltage control method to provide an almost sinusoidal voltage to the motor windings. II. PROPOSED FILTER TOPOLOGY When the PMSM is controlled by FOC with hysteresis current controllers, the motor line currents are controlled to oscillate within a predefined hysteresis band. Fig. 1 shows the current waveform and the associated inverter output voltage switching. In the figure, the inverter changes state at the end of a sampling period only when the actual line current increases or decreases beyond the hysteresis band, which results in a high ripple current full of harmonic components. To reduce the severity of this ripple, two methods can be mentioned. The first method is to reduce the sampling period, which implies very fast switching elements. The second method is to affect the voltage provided to the motor terminals in such a way as to almost follow a sinusoidal reference guide. The last method is adopted here. On that account, the AF topology, as Fig. 1. Current waveform and associated inverter voltage switching of the FOC equipped with hysteresis current controllers. Fig. 2. Basic structure of the proposed filter topology. shown in Fig. 2, is used to affect the inverter voltage waveform to follow the required signal voltage as in Fig. 1. Fig. 2 shows a schematic of the basic structure of the pro- posed filter topology, including the AF, coupling transformer, RLC filters, and block diagram of the AF control circuit. In Fig. 2, Vsig is the desired voltage to be injected in order to obtain a sinusoidal voltage at the motor terminals, and VAF is the measured output voltage of the AF. VAF is subtracted from Vsig and passed to the hysteresis controller in order to generate the required switching signal to the AF. The AF storage capacitor CF , which operates as the voltage source, should carefully be selected to hold up to the motor line voltage. The smoothing inductance LF should be large enough to obtain an almost sinusoidal voltage at the motor terminals. The reference sinusoidal voltage V ∗, which should be in phase with the main inverter output voltage Vinv, is calculated using the information of the motor variables. The proposed filter topology consists of three parts, i.e., one is the voltage reference circuit based on the space vector GULEZ et al.: TORQUE RIPPLE AND EMI NOISE MINIMIZATION IN PMSM USING AF TOPOLOGY AND FOC 253 calculation, another is the AF part, and the other is the coupling part, which consists of a 1 : 1 transformer and two RLC filters. In the coming sections, first, the operating principle of the voltage reference control circuit will be explained, then, the two other parts will follow. A. Voltage Reference Signal Generator The effectiveness of the AF is mainly defined by the algo- rithm that is used to generate the reference signals required by the control system. These reference signals must allow current and voltage compensation with minimum time delay. In this paper, the method used to generate the voltage reference signals is related to the control algorithm of the motor, which uses the motor model in the rotor d–q reference frame and the rotor FOC principles with monitored rotor position/speed and monitored phase currents. The motor model in this synchronously rotating reference frame is given by[ vsd vsq ] = [ R+ pLsd −PωrLsq PωrLsd R+ pLsq ] [ isd isq ] + [ 0 eB ] (1) Te = 3 2 P (ψF isq + (Lsd − Lsq)isdisq)) (2) where vsd, vsq d-axis and q-axis stator voltages; isd, isq d-axis and q-axis stator currents; R stator winding resistance; Lsd, Lsq d-axis and q-axis stator inductances; p = d/dt, differential operator; P number of pole pairs of the motor; ωr rotor speed; ψF rotor permanent magnetic flux; eB = PωrψF , generated back EMF due to ψF ; Te generated electromagnetic torque. Under base speed operation, the speed or torque control can be achieved by forcing the stator current component isd to be zero while controlling the isq component to be directly proportional to the motor torque Te as Te = 3 2 PψF isq. (3) The instantaneous q-axis current can be extracted from (3). Hence, by setting isd to zero, the instantaneous d- and q-axis voltages can be calculated from (1) as Vsd = − PωrLsqisq (4) Vsq =Risq + pLsqisq + PωrψF . (5) Once the values of the d- and q-axis voltage components are obtained, the Park and Clarke transformation can be used to obtain the reference sinusoidal voltages as   v ∗ a v∗b v∗c   = K   1 0−1/2 √3/2 −1/2 −√3/2   [ cos θ − sin θ sin θ cos θ ] [ Vsd Vsq ] (6) where K is the transformation constant, and θ is the rotor position. Fig. 3. Simplified power circuit of the proposed filter topology. Fig. 4. Coupling circuit between the AF and the main inverter on one side and the PMSM on the other side. B. AF Compensation Circuit Fig. 3 shows a simplified power circuit of the proposed topology (the passiveRCL filters are not shown). In this circuit, Vdc is the voltage of the main inverter circuit, and V ±CF is the equivalent compensated voltage source of the AF. In order to generate the required compensation voltages that follow the voltage signal vsig, bearing in mind that the main inverter changes switching state only when the line current violates the condition of the hysteresis band and that the capacitor voltage polarity cannot abruptly change, the switches sw1 and sw2 are controlled within each consecutive voltage switching of the main inverter to keep the motor winding voltages within the acceptable hysteresis band. The motor line current im is controlled within the motor main control circuit with hysteresis current controller to provide the required load torque; therefore, two hysteresis controller systems (i.e., one for voltage and the other for current) are independently working to supply the motor with an almost sinusoidal voltage. In Fig. 3, when the switching signal (e.g., 100) is sent to the main inverter, i.e., phase a is active high while phases b and c are active low, then, following the path of the current im in Fig. 3, the voltage provided to the motor terminal can be expressed as Vs = 2 3 ( Vdc − V ±CF − 3 2 LF dim dt ) . (7) The limit values of the inductor LF and capacitor CF can be determined as follows. 254 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 1, JANUARY 2008 TABLE I MOTOR PARAMETERS TABLE II L-TYPE FILTER PARAMETERS Fig. 5. Motor phase voltage before and after applying the AF. During a sampling period Ts, the change in capacitor voltage can be calculated as ∆VCF = 1 CF Ts∫ 0 imdt. (8) So if the maximum capacitor voltage change is determined as Vdc, the minimum capacitor value can be calculated as CF ≥ ∣∣∣∣∣ ∫ (n+1)Ts nTs imdt V dc ∣∣∣∣∣ = ∣∣∣∣Ts • imavVdc ∣∣∣∣ (9) where imav is the maximum of the average current change that can occur per sample period. The limit values of the smoothing inductance LF can be expressed as 1 (2πfsw)2CF < LF ≤ ∣∣∣∣∣ VLFmax 3 2 max ( dim dt ) ∣∣∣∣∣ (10) where the lower limit is determined by selecting the resonance frequency of the combination CFLF to be less than the inverter switching frequency fsw to guarantee reduced switching fre- quency harmonics. The upper limit is calculated by determining the maximum voltage drop across the inductors VLFmax and the maximum current change per sampling period dim/dt. Fig. 6. Injected voltage from AF. Fig. 7. Motor lines current before and after applying the AF. Fig. 8. Motor torque before and after applying the AF. C. Coupling The coupling between the main inverter circuit and the AF circuit is achieved through a 1 : 1 transformer, and to attenuate the higher-frequency EMI noises, the LCR filters are used at the transformer primary and secondary windings, as suggested in Fig. 4. The important point here is that the resonance that may arise between the capacitor C1 and transformer primary winding and between the capacitor C2 and motor inductance winding should be avoided when selecting capacitor values. At a selected cutoff frequency, the currents iCR1 and iCR2 derived by the RLC filters are given by iCR1 = zT zT + √ R1 + 1/sC1 im1 iCR2 = zPMSM zPMSM + √ R2 + 1/sC2 im2 (11) where zT and zPMSM are as defined in Fig. 4. GULEZ et al.: TORQUE RIPPLE AND EMI NOISE MINIMIZATION IN PMSM USING AF TOPOLOGY AND FOC 255 Fig. 9. Rotor speed before and after applying AF. Fig. 10. Phase a current (upper) and its spectrum (lower) before connecting the AF. At the selected cutoff frequency, these currents should be large compared to im1 (which is drawn by the transformer) and/or im (which is drawn by the motor). On the other hand, at the operating frequency, these currents should be very small compared to im1 and im. Another point in the selection of the RLC parameters is that the filter inductors are essentially shorted at the line frequency while the capacitors are open circuit, and for the EMI noise frequencies, the inductors are essentially open circuit while the capacitors are essentially shorted; thus, a considerable amount of EMI noises will pass through the filter resistors to the earth and cause a frequency- dependent voltage drop across the inductors that in turn will help in smoothing the voltage waveform supplying the motor. Fig. 11. Phase a current (upper) and its spectrum (lower) after connecting the AF. III. SIMULATIONS AND RESULTS To simulate the performance of the proposed filter topology, Matlab/Simulink was used. The effectiveness of the filter topol- ogy was shown by providing the filter into operation while the motor is running. The PMSM is star connected with earth return. The motor parameters are shown in Table I, while the passive filter param- eters are shown in Table II. The AF capacitor that was used is 200 µF, and its inductors are 200 mH. A. Motor Performance The simulation results with 100-µs sampling time are shown in Figs. 5–13. Fig. 5 in particular shows the phase voltage provided to the motor terminals. Observing the change of the waveform after switching on the AF (at time = 0.15 s) into the circuit, it is clear that the phase voltage approaches a sinusoidal waveform. Fig. 6 shows the injected voltage from the AF. A better waveform can be obtained by increasing the AF inductance LF . However, the cost and size of the AF will increase, so an acceptable inductance value can be selected to achieve less than 2% of the total harmonic distortion (THD). The motor performances before and after applying the AF are shown in Figs. 7–9. In Fig. 7, the motor line currents show considerable reduction in noise and harmonic components after applying the AF, which is reflected in a smoother current waveform. 256 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 1, JANUARY 2008 Fig. 12. EMI noise level before connecting the AF. Fig. 13. EMI noise level after connecting the AF. The torque response in Fig. 8 shows a dramatic drop in torque ripple from 3.2 to 0.2 N ·m after applying the AF, which will result in reduced motor mechanical vibration and acoustic noise. This reduction is also reflected in a smoother speed response, as shown in Fig. 9. B. Harmonics and EMI Noise Reduction The status of the line current harmonics and the EMI noise before and after connecting the AF are shown in Figs. 10–13. In Fig. 10, the spectrum of the line current before connecting the AF shows that disastrous harmonics currents with THD of ∼15% have been widely distributed with a dominant harmonics amplitude of ∼16% in the range of thirtieth to fiftieth harmonic order. After connecting the AF, the THD is effectively reduced to less than 1.5% with dominant harmonics amplitude of ∼1% in the range greater than the eight harmonic order, as shown in Fig. 11. The EMI noise level before connecting the AF in Fig. 12 shows