27637_ftp

4.2 mm, H1 ¼ 1.1 mm, H2 ¼ H3 ¼ 1.36 mm, H4 ¼ 1.36 mm, L5 ¼ 6 mm, L6 ¼ 6.5 mm, L7 ¼ L11 ¼7.4 mm, L8 ¼ L10 ¼ 10 mm, L9 ¼ 10.6 mm, and L12 ¼ 12.5 mm. The experimental cir- cuit is fabricated on a substrate with relative dielectric constant of 2.65 and the thickness of 1 mm by standard PCB process. The manufactured prototype of the proposed power divider is shown in Figure 4. The size of the proposed power divider is about 32 � 45 mm2, which does not include the size of 50-X feeding line and the length of the transition between the SIW and the microstrip. The measured insertion losses and the input return loss for the two-way equal SIW power divider can be observed in Figure 5, where they are compared with full-wave simulated results provided by the commercial software Ansoft HFSS. According to the measurements, the transmission response for the proposed power divider falls to at 3.48 GHz and at 5.2 GHz and the power divider has an insertion loss level of 3.7 6 0.5 dB, which includes 3.01-dB inherent insertion loss of two-way equal power divider. With regard to the input return loss, the power divider has a minimum value of 10.6 dB throughout the pass band. The out-of-band rejection is larger than 40 dB at 2.35 GHz and at 6.12 GHz, respectively. 4. CONCLUSION In this article, a compact two-way equal SIW power divider with sharp out-of-band rejection has been presented. Experimen- tal power divider has been built and measured. Good agreements are observed between the measured results and the simulations. This power divider is very suitable for application in low cost and compact size microwave and millimeter-wave planar circuits by traditional PCB technology. ACKNOWLEDGMENT This present work was supported by the Fundamental Research Funds for the Central University (No. CDJRC11160001). REFERENCES 1. J.-Y. Shao, S.-C. Huang, and Y.-H. Pang, Wilkinson power divider incorporating quasi-elliptic filters for improved out-of-band rejec- tion, Electron Lett 47 (2011), 1288–1289. 2. J. Wang, J. Ni, and Y.-X. Guo, Miniaturized microstrip Wilkinson power divider with harmonic suppression, IEEE Microwave Wirel Compon Lett 19 (2009), 440–442. 3. T. Skaik, M. Lancaster, and F. Huang, Coupled-resonator 3-dB power divider, In: Proceedings of IET seminar on passive RF and microwave components, Birmingham, UK, 2010, 21–22. 4. T. Skaik, M. Lancaster, and F. Huang, Synthesis of multiple output coupled resonator circuits using coupling matrix optimization, IET Microwave Antennas Propag 5 (2011), 1081–1088. 5. X. Zhang, J. Xu, and Z. Yu, C-band half mode substrate integrated waveguide (HMSIW) filter, Microwave Opt Tech Lett 50 (2008), 275–277. 6. Y.Q. Wang, W. Hong, Y.D. Dong, B. Liu, H.J. Tang, J.X. Chen, and K. Wu, Half mode substrate integrated waveguide (HMSIW) band- pass filter, IEEE Microwave Wirel Compon Lett 17 (2007), 265–267. VC 2013 Wiley Periodicals, Inc. BROADBAND BANDPASS FILTER USING PARALLEL-COUPLED MICROSTRIP LINE AND COUPLED OPEN STUB RESONATOR Jayaseelan Marimuthu, Amin Abbosh, and Bassem Henin School of ITEE, The University of Queensland, St Lucia QLD 4072, Australia; Corresponding author: j.marimuthu@uq.edu.au Received 31 October 2012 ABSTRACT: A novel broadband bandpass filter with multiple resonant modes based on parallel-coupled microstrip line is proposed. The required cutoff frequency and out-of-band performance are achieved by placing L-shaped capacitive cross-coupling open stubs at the middle resonator with appropriate dimensions. A compact broadband bandpass filter of dimensions 18 � 29 mm2 is fabricated and tested for performance confirmation. The proposed filter demonstrates a wide bandwidth (from 3 to 7 GHz), as well as excellent out-of-band performance with more than 25-dB rejection up to more than 12 GHz and sharp upper cutoff frequency due to the proper location of two transmission zeros. VC 2013 Wiley Periodicals, Inc. Microwave Opt Technol Lett 55:1640–1644, 2013; View this article online at wileyonlinelibrary.com. DOI 10.1002/mop.27637 Key words: broadband bandpass filter; parallel-coupled microstrip line; microstrip filter 1. INTRODUCTION Compact broadband filters which are compatible with printed circuit boards are highly demanded in most of the modern com- munication systems. The filter size is usually constrained by the number of resonators and size of the resonator structures used in the design, whereas the filter bandwidth is mainly limited by the achievable maximum coupling between its resonators. Various compact resonator structures were introduced in literature [1–4]. Parallel-coupled microstrip line (PCML) structure has been used as a coupling component in the design of bandpass filters [5, 6]. To achieve a broadband bandpass filter, a high-coupling PCML structure constructed from narrow width parallel microstrip lines with narrow gap in-between. Various broadband and ultrawide- band (UWB) bandpass filters using PCML structure were reported in literature [7–11]. A ground plane aperture technique for PCML structure was proposed and developed to enhance the tight coupling over a certain frequency range [7]. Multipole broadband microstrip bandpass filter was realized by attaching a single-line high-im- pedance resonator of uniform line section between two PCML sections with backside aperture. The overall proposed design required a ground plane aperture and a pair of capacitive open- ended stubs. On the basis of impedance steps and PCML as Figure 5 The measured results compared with simulated results for the proposed power divider from 1 to 8 GHz J_ID: ZS4 Customer A_ID: MOP-12-1255 Cadmus Art: MOP27637 Date: 16-April-13 Stage: Page: 1640 ID: sakthivelm I Black Lining: [ON] I Time: 00:31 I Path: N:/3b2/MOP#/Vol00000/130117/APPFile/JW-MOP#130117 1640 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 7, July 2013 DOI 10.1002/mop inverter circuits, various designs of broadband bandpass filters were presented in Ref. 8. The proposed design uses suspended stripline of PCML structure to reduce the spurious response and enhance the coupling factor. A novel microstrip-line UWB bandpass filter with compact size was proposed in Ref. 9 by forming a multiple-mode resonator and introducing quarter- wavelength PCML in the input and output ports. It has been demonstrated that a feeding network with low- characteristic impedance is able to improve both the coupling factor and operating bandwidth of PCML structure [10]. Similar to the proposed filter in Ref. 7, a broadband PCML structure was designed in Ref. 10 without having a backside aperture and a pair of capacitive open-ended stubs. Based on the structure in Ref. 10, a compact UWB bandpass filter with four resonant modes was proposed in Ref. 11. The proposed filter was based Figure 1 (a) Circuit model of the initial proposed design. (b) Layout of the initial proposed design Figure 2 Full-wave simulated response of the design for various val- ues of Scle based on Figure 1(b). (a) S11 and (b) S21. [Color figure can be viewed in the online issue, which is available at wileyonlinelibrary.com] Figure 3 (a) Circuit model of the proposed filter with a pair of L- shaped parallel open stub. (b) Layout of the proposed filter Figure 4 Full-wave simulated response of the design with parallel open stub for various values of Los with Wos ¼ 2.2 mm. (a) S11 and (b) S21. [Color figure can be viewed in the online issue, which is available at wileyonlinelibrary.com] J_ID: ZS4 Customer A_ID: MOP-12-1255 Cadmus Art: MOP27637 Date: 16-April-13 Stage: Page: 1641 ID: sakthivelm I Black Lining: [ON] I Time: 00:31 I Path: N:/3b2/MOP#/Vol00000/130117/APPFile/JW-MOP#130117 DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 7, July 2013 1641 on a PCML structure with L-shaped low impedance feeding net- work and C-shaped low impedance middle resonators. In this article, a compact broadband PCML structure similar to Refs. 10 and 11 is designed. The proposed design uses a sin- gle-line low impedance resonator of specific length and width with a pair of cross-coupled L-shaped open-ended stubs attached between the two PCML sections. Moreover, a low impedance feeding network is introduced at the end of the structure. The cross-coupled L-shaped open-ended stubs at the middle resona- tor are used to improve the performance of the filter. It is also used to improve the upper cutoff frequency based on creating the transmission zero of the design in the required position; a property that was not realized in Refs. 10 and 11. The position of the PCML feeding at the middle resonator embedded with open-ended stub forms quarter-wavelength open-ended stubs. Those stubs produce a standing wave at a certain frequency to initiate a transmission zero. In the higher stopband, the proposed filter achieves a sharp cutoff response due to the transmission zero of the proposed resonators. That response is based on the length of the open-ended stubs and the cross-coupling between the stubs. The structure of the filter is optimized using an ADS momentum. The design is accomplished using the substrate Rogers RT6006 with dielectric constant of 6.15 and thickness of 1.27 mm. The filter is then fabricated and tested to validate the proposed method. 2. FILTER DESIGN The initial design of the filter is based on the structure depicted in Figure 1. In this case, a simple PCML broadband bandpass filter is designed using quarter wavelength (k/4) resonators. The initial filter design based on Ref. 10 uses a rectangular middle resonator with Wrmr ¼ 2.6 mm (Zo2 ¼ 41 X) and Lrmr ¼ 7 mm (�k/4 at 4.85 GHz). In addition to that, it uses a pair of tight PCML structure with Lcl ¼ 7.3 mm (�k/4 at 4.85 GHz), Wcl ¼ 0.4 mm, and Scl ¼ 0.2 mm with odd impedance Zoo ¼ 51.5 X and even impedance Zoe ¼ 137.5 X. The structure is connected to the middle resonator at Lp ¼ 0.8 mm with feeding network Lfn ¼ 3.5 mm and Wfn ¼ 2 mm (Zo1 ¼ 48 X). A cross-coupling capacitor C1, as shown in Figure 1(a), is then introduced between the PCML structure and the middle resonator. The ca- pacitor is added by creating a gap of Scle between one of the strips of the PCML structure, which is connected to the feeding network and the middle resonator. This capacitance is able to produce multiple transmission zeros within the design. The design based on Figure 1(b) with the above configuration is simulated using ADS momentum full-wave simulator. The simu- lation runs using the substrate Rogers RT6006 with dielectric constant of 6.15 and thickness of 1.27 mm for various Scle. Figure 2 shows the simulated transmission coefficient S21 and reflection coefficient S11 for the proposed design for various Scle based on Figure 1(b). The simulated results for S11 show multiple resonances at various frequencies. The first two resonances f1 � 3.0 and f2 � 4.5 GHz can be considered as the required band for the design while the higher resonances (fh1 and fh) can be classified as higher harmonics. The simulated results for Ss1 show transmission zeros at �9.9 and �11.0 GHz. When Scle ¼ 2.0 mm, the transmission coefficient is S21 � �1.9 dB and reflection coefficient S11 � �5.0 dB with operating band from 2.7 to 5.5 GHz. When Scle ¼ 1.0 mm, the transmission coefficient is S21 � �0.9 dB and reflection coefficient S11 � �7.0 dB with operating band from 2.92 to 6.2 GHz. For the two cases Scle ¼ 2.0 mm and Scle ¼ 1.0 mm, as Scle/Wcl 1, the effect of the capacitive cross-coupling C1 is negligible. The design shows weak coupling of the PCML structure which leads TABLE 1 The Summary of the Design Based on Figure 4 for Various Stub Lengths Los (mm) Cos (pF) Resonant Frequencies (f1 and f2) Harmonic Frequency (fh) Transmission Zero Further Harmonics 0 0.0 3.50 GHz, 5.76 GHz 9.35 GHz 9.52 GHz, 11.0 GHz No 1 0.186 3.52 GHz, 4.70 GHz 8.70 GHz 9.18 GHz, 11.24 GHz No 2 0.397 3.55 GHz, 4.44 GHz 7.85 GHz 8.73 GHz No 3 0.679 3.54 GHz, 4.23 GHz 6.90 GHz No 9.65 GHz, 11.30 GHz 4 1.140 3.53 GHz, 4.12 GHz 5.84 GHz No 8.55 GHz, 10.70 GHz TABLE 2 Summary of the Design Based on Figure 5 for Various Extended Stub Length Lf and Gap Sg Capacitance Due to Sg Lf (mm) Capacitance of (LosþLf) Stub Cosf, (pF) Sg (mm) Cp (pF) [6] Cg (pF) [6] Sg/Wf 0.0 0.0 2.6 0 0 >>1 0.8 0.908 1.0 35.35 24.47 0.625 1.2 1.09 0.2 8.56 76.56 0.125 Figure 5 Full-wave simulated response of the design with parallel open stub with extended length to form capacitive coupling for various values of Lf with Wf ¼ 1.6 mm. [Color figure can be viewed in the online issue, which is available at wileyonlinelibrary.com] J_ID: ZS4 Customer A_ID: MOP-12-1255 Cadmus Art: MOP27637 Date: 16-April-13 Stage: Page: 1642 ID: sakthivelm I Black Lining: [ON] I Time: 00:32 I Path: N:/3b2/MOP#/Vol00000/130117/APPFile/JW-MOP#130117 1642 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 7, July 2013 DOI 10.1002/mop to poor passband and stopband performances with poor upper cut-off frequency. On the other hand, when Scle ¼ 0.2 mm (Scle/Wcl 1), the design shows a good passband and stopband performance with transmission coefficient S21 � �0.5 dB and reflection coefficient S11 � �11 dB with operating band from 3.0 to 6.6 GHz. It also shows a sharp upper cut-off frequency with resonance frequen- cies at f1 ¼ 3.5 GHz, f2 ¼ 5.76 GHz, and harmonic frequency at fh1 ¼ 9.35 GHz. The design shows excellent capacitive cross- coupling effects C1 and strong coupling on the PCML structure which adds transmission zero at 9.52 and 11.0 GHz. The circuit modal and the layout of the new proposed design are shown in Figure 3. The design is developed from the origi- nal structure in three steps. First, the design is improved by placing a parallel open stub (Z03) of width Wos and length Los at both ends of the middle resonator as shown in Figure 3. The parallel open stub is designed by using width Wos ¼ 2.2 mm (Z03 ¼ 45.5 X) and length Los < kg/4. Because Los < kg/4, the input admittance of the stub is capacitive. Based on above con- figuration, the design was simulated using ADS. Figure 4 shows the transmission coefficient and reflection coefficient of the design for various lengths Los of the parallel open stubs. Table 1 shows the equivalent capacitance Cos [5] for the var- ious stub length Los at 4.85 GHz. It shows that as the stub length increases the equivalent capacitance Cos increases and the resonance frequencies within the operating band and the har- monic frequencies moves toward lower values. The first har- monic frequency is initially at 9.35 GHz for Los ¼ 0 mm and moves toward the operating band with increasing Los. When Los ¼ 4 mm, that frequency moves to 5.84 GHz and becomes part of the operating band that has effectively three resonant frequen- cies. As the stub length increases, no significant transmission zero presents and multiple harmonics appear that affect the filter performance in the stopband significantly. Based on Table 1 and ADS optimization, the optimum stub length required to produce a transmission zero is Los ¼ 2.9 mm. The equivalent capacitance of the stub is Cos ¼ 0.645 pF at 4.85 GHz. This stub is used as a parallel open stub at both ends of the middle resonator to design a broadband bandpass filter. The transmission coefficient of S21 � �4.1 dB and reflection coefficient of S11 � �2.4 dB are achieved throughout the oper- ating band 3.1–7.1 GHz with sharp upper cut-off frequency. The design has three resonant frequencies at 3.6, 4.2, 7.0 GHz, and a transmission zero at 8.4 GHz. The design is further improved by placing an extended stub of high impedance Zo4 ¼ 54.5 X with a width of Wf ¼ 1.6 mm and a length of Lf as shown in Figure 3. The added stub form a capacitive cross-coupling Cg between the edge of both stubs and a shunt capacitance Cp as listed in Table 2. The design is simu- lated for various configurations of Lf and Sg. Figure 5 shows the insertion and reflection coefficient of the filter without extended stub and with extended stub of length Lf ¼ 0.8 and 1.2 mm. For Lf ¼ 0.8 mm with a gap between the two edge of open stubs Sg ¼ 1.0 mm, the transmission coefficient is S21 � �1.6 dB, and the reflection coefficient is S11 � �5.7 dB throughout a pass- band of 3.1–6.7 GHz with a sharp upper cut-off frequency. The filter has three resonant frequencies at 3.56, 4.16, 6.59 GHz, and a transmission zero at 8.16 GHz. By extending the stub length to Lf ¼ 1.2 mm with Sg ¼ 0.2 mm, the cross-coupling capacitance becomes Cg ¼ 76.56 pF. This value is very high compared to Cosf ¼ 1.09 pF (Table 1) and Cp ¼ 8.56 pF. The high capacitance value improves the overall performance of the filter in the passband and stopband with sharp upper cut-off frequency with excellent transmission zero. The design with the above specifications has excellent transmission coefficient S21 � �0.3 dB and reflection coefficient S11 � �13.7 dB throughout the passband from 3.1 to 6.3 GHz. It has three resonant frequencies at 3.5, 4.2, 5.96 GHz, and a transmission zero at 7.3 GHz with S21¼�50 dB. The filter Figure 7 Photograph of the fabricated filter. [Color figure can be viewed in the online issue, which is available at wileyonlinelibrary.com] TABLE 3 The Summary of the Optimized Design Parameters Based on Figure 3(b) Parameter Value (mm) Wrmr 2.6 (Zo2¼ 41 X) Lrmr 7.0 (�k/4 at 4.85 GHz) LOS 2.9 WOS 2.2 (Zo3¼ 45.5 X) Lf 1.2 Wf 1.6 (Zo4¼ 54.5 X) Sg 0.2 Lcl 7.3 (�k/4 at 4.85 GHz) Wcl 0.4 Scl 0.2 Scle 0.2 Lp 2.8 Lfn 3.5 Wfn 2.0 (Zo1¼ 48 X) Figure 6 Full-wave simulated response of the proposed design for various values of Lp. [Color figure can be viewed in the online issue, which is available at wileyonlinelibrary.com] J_ID: ZS4 Customer A_ID: MOP-12-1255 Cadmus Art: MOP27637 Date: 16-April-13 Stage: Page: 1643 ID: sakthivelm I Black Lining: [ON] I Time: 00:32 I Path: N:/3b2/MOP#/Vol00000/130117/APPFile/JW-MOP#130117 DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 7, July 2013 1643 shows excellent broad stopband from 6.3 to 12.6 GHz with S21 � �25 dB as shown in Figure 5. The design is further improved by varying the position Lp of the PCML structure on the middle resonator. Figure 6 shows the transmission coefficient and reflection coefficient of the filter for different values of Lp. As the Lp increases from its original value 0.8 to 2.8 mm, the passband bandwidth increases with highly sharper upper cut-off frequency and excellent transmis- sion zero. The transmission zero is effectively shifted to higher frequency which leads to increase in the bandwidth. That shift in the transmission zero is due to the decrease in the effective length of the distance between the open-ended stubs and the feeding PCML. In the case of Lp ¼ 0.8 mm, the transmission zero is at 7.1 GHz because the overall length of the open-ended stubs is quarter wavelength at 7.1 GHz. Similarly when Lp ¼ 2.8 mm, the transmission zero is at 8.1 GHz as the overall length of the open-ended stubs is quarter wavelength at 8.1GHz. The design is then optimized by placing a notch at both inner corners of the middle resonator to reduce the losses due to sharp edges at 4.85 GHz. The added notches, as shown in Figure 3(b), improve the performance of the passband for both the transmis- sion coefficient and the reflection coefficient as shown in Figure 6. The optimized filter has a broad bandwidth more than 4 GHz centered at 4.85 GHz. The optimized design shows excellent passband and stopband performance of transmission coefficient and reflection coefficient with a sharp transmission zero at 8.0 GHz with S21 ¼ �63.44 dB as shown in Figure 6. 3. EXPERIMENTAL VERIFICATION Based on the above simulations, the dimensions of the filter with the best performance are determined as shown in Table 3. The simulated results of the transmission coefficie